High isolation power combiner/splitter and coupler

ABSTRACT

A power combiner has at least two uncommon ports and at least one common port. Isolation between the uncommon ports over a broad band is achieved through a lossy band response circuit having a phase and amplitude response effective to compensate for changes in phase and amplitude between the uncommon ports with change in frequency of an input signal. The lossy band response circuit may have a resistance approximating a resistance effective to isolate the uncommon ports over a bandwidth at a center frequency. A coupler may likewise increase the band for which an input port is isolated from the isolated port by coupling a lossy circuit between the input port and isolated port. The lossy circuit may be embodied as a lossy band response circuit.

RELATED APPLICATION

This patent application is a divisional (continuation) of co-pendingU.S. patent application Ser. No. 14/854,346, filed Sep. 15, 2015, whichwill issue Jul. 18, 2017 as U.S. Pat. No. 9,712,131, and which is herebyincorporated herein by reference.

BACKGROUND Field of Invention

This invention relates to high isolation power combiners/splitters andcouplers, specifically to such combiners/splitters and couplers that areused in RF and microwave circuits.

Description of Prior Art

Special circuits are sometimes required to combine two or more highfrequency RF or microwave signals. A commonly used circuit that canaccomplish this is called a power combiner. An N-way combiner has onecommon port and N uncommon ports. The ports are places in the circuitthat allow intentional connection to other external circuits. Ingeneral, a good power combiner should sum N input signals together withminimal signal energy or power loss. It should also keep the signalsthat are being summed together isolated from each other at the uncommoninputs. High uncommon port to uncommon port isolation can help to avoidimpedance mismatches and signal degradation, and helps to minimizestress to circuitry connected to the uncommon ports.

A power combiner can sometimes be used for the purpose of splitting asingle signal into two or more signals. When used in this configuration,the circuit is typically called a power splitter or power divider.

Additionally, sometimes special circuits are used to split signals orsample signals. Couplers, which may be thought of as a special case of apower splitter, are used to couple or sample some RF or microwave signalenergy that may be used for various purposes. Often, it is desirablethat a coupler have a coupled port and an isolated port, where theisolated port has little or none of the sampled signal, and the coupledport have a specified portion of the sampled signal. Some couplers areused to determine the direction of the sampled signal, whether it flowsinto the input port, or out of the input port. In these cases, thecoupler directivity becomes an important parameter. Directivity is afunction of the coupler isolation, again making it desirable for thecoupler to have high isolation. In these cases, the coupler is oftenreferred to as a Directional Coupler.

Power Combiners/Splittters

Before the 1960's, power combiner/splitter circuits tried to minimizeinsertion losses, but did not have significant port to port isolation.Circuits such as the hybrid T and quarter wave matched splitters werecommon. (U.S. Pat. No. 2,877,427—Butler and 2963664—Yeagly). Othercombiner/splitter designs, (U.S. Pat. No. 3,529,265—Podell and ‘A NewN-way Power Divider/Combiner Suitable for High Power Applications, Proc.Of 1975, IEEE MTT Seminar’—Gysel) demonstrate designs that exhibitisolation between uncommon ports but also are less sensitive to loadmismatches and can accommodate high power signals.

U.S. Pat. No. 3,091,743, issued to Wilkinson, disclosed a powercombiner/splitter that achieved relatively high port to port isolationbetween uncommon ports. The isolation was achieved by providingadditional signal paths between the uncommon ports through resistors.The portion of a signal traveling through a resistor path hadapproximately the same amplitude as the signal that traveled to thecommon port and back to an adjacent uncommon port, but it wasapproximately 180 degrees out of phase, at the center frequency ofdesign, having a zero degree phase shift at the center frequency ofdesign. This created relatively high isolation between uncommon ports,due to the canceling effect of this circuit topology. For a singlesection design, however, port to port isolation greater than 40 or 50 dBbetween the uncommon ports, occurred only in a relatively smallfrequency bandwidth, on the order of 1 percent. This high isolationoccurred at the center frequency of the design.

Cohn published a paper in 1968 (IEEE transactions on Microwave Theoryand Techniques, Vol. MTT-16, No. 2, February 1968) where he introduced amultiple section power combiner/splitter. Cohn found that by addingadditional quarter wave transmission line sections, plus an additionalisolation resistor for each section, that high isolation betweenuncommon ports may be achieved over a broader bandwidth than a singlesection. A two section design can achieve up to 50 dB of isolationbetween uncommon ports over a 20 percent bandwidth, and up to 70 dB in a5 percent bandwidth. While this design offered a better solution to manysystems, this approach has the drawback of requiring additional circuitsize due to additional transmission line sections. It also suffers fromincreased insertion loss, since the signal must travel through longertransmission lines. For many circuits, the relatively large size of amultisection power combiner cannot be tolerated. Additional insertionloss as well, is almost always a drawback in power combiners.

Since 1968, further advances of power combiners/splitters have beenaccomplished. Some designs exhibit improved power handling capability;others demonstrate decreased circuit size and others improvemanufacturability. Some designs incorporate these improvementsseverally. Advances have also improved or modified electrical parametersof power combiners/splitters such as increased bandwidth, improved inputand output VSWR and slightly lower insertion loss. However, to datethere has been little improvement on increasing the magnitude of thein-band isolation between uncommon ports.

U.S. Pat. No. 5,489,880, issued to Swarup, discloses a powercombiner/splitter that has increased port to port isolation forout-of-band signals. This is achieved by putting band pass filters atthe uncommon ports. While this improves the out-of-band port to portisolation, the in-band isolation is not significantly improved. Thisdesign also suffers from increased insertion loss over a typical powercombiner/splitter due to the signal going through a band pass filter aswell as the combiner/splitter circuit.

Some systems require an uncommon port to port isolation that is higherthan existing power combiner/splitters can provide. Placing circulatorsor isolators at uncommon ports, one can achieve isolations greater than50 dB over relatively large bandwidths. These products, however, arerelatively costly and exhibit non-linear characteristics at high inputsignal levels. They also increase the signal path insertion loss.Furthermore, they are not reciprocal circuits, and cannot be used assplitters without re-orienting the circulators or isolators.

SUMMARY OF THE INVENTION—OR—OBJECTS AND ADVANTAGES

Accordingly, it is an object of the present invention to provide a novelN-way power combiner/splitter and Coupler having the followingadvantages:

-   -   1) Ultra-high isolation of up to 50 dB between uncommon ports of        up to 20% bandwidth or more for a single section power        combiner/splitter.    -   2) Ultra-high isolation of up to 70 dB between uncommon ports of        up to 5% bandwidth or more for a single section power        combiner/splitter.    -   3) Does not have significant degradation of insertion loss over        a standard power combiner/splitter using only resistive elements        for isolation.    -   4) May be constructed using either distributed and        lumped-element design techniques or only lumped-element design        techniques.    -   5) Does not exhibit non-linear characteristics for high input        signals.    -   6) Is a reciprocal circuit in that the same configuration can be        used for power combining as well as power splitting.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a prior art circuit that is a 2-way Wilkinson powercombiner/splitter using distributed transmission lines and a lumpedelement isolation resistor.

FIG. 2 is a schematic circuit diagram of one embodiment of a 2-waymodified Wilkinson power combiner/splitter of the present inventionusing distributed transmission lines and lumped element isolationcomponents in a lossy band response configuration connected betweenuncommon ports.

FIG. 3 is a graph showing comparison plots of the uncommon port touncommon port isolation of a single section Wilkinson powercombiner/splitter demonstrating the prior art isolation performance andtwo implementations of the present invention circuits, which have moreuncommon port to uncommon port isolation over a wider bandwidth.

FIG. 4 is a schematic circuit diagram showing some of the multipleembodiments possible of the present invention.

FIG. 5 illustrates a prior art circuit that is a 3-way powercombiner/splitter using distributed transmission lines and lumpedelement isolation resistors, where the lumped element isolationresistors connect between uncommon ports.

FIG. 6 is a schematic circuit diagram of one embodiment of a 3-way powercombiner/splitter of the present invention using distributedtransmission lines and lumped element isolation components in a lossyband response circuit configuration, connected between uncommon ports.

FIG. 7 is a graph showing comparison plots of the uncommon port touncommon port isolation of a 3 way power combiner/splitter demonstratingthe prior art isolation performance and the advantages of twoimplementations of the present invention circuits, which have moreuncommon port to uncommon port isolation over a wider bandwidth.

FIG. 8 illustrates a prior art circuit that is a 3-way Wilkinson powercombiner/splitter using distributed transmission lines and lumpedelement isolation resistors connected to a common point in the circuit.

FIG. 9 is a schematic circuit diagram of one embodiment of a 3-waymodified Wilkinson power combiner/splitter of the present inventionusing distributed transmission lines and lumped element isolationcomponents in a lossy band response circuit configuration, connected toa common point in the circuit.

FIG. 10 is a graph of comparison plots of the uncommon port to uncommonport isolation of a 3 way Wilkinson power combiner/splitterdemonstrating the prior art isolation performance and the advantages oftwo implementations of the present invention circuits, which have moreuncommon port to uncommon port isolation over a wider bandwidth.

FIG. 11 illustrates a prior art circuit that is a 2-way lumped elementpower combiner/splitter using lumped elements having a high passconfiguration and a lumped element isolation resistor connected betweenuncommon ports.

FIG. 12 is a schematic circuit diagram of one embodiment of a 2-waymodified lumped element power combiner/splitter of the present inventionusing lumped elements having a high pass configuration and lumpedelement isolation components in a lossy band response circuitconfiguration connected between uncommon ports.

FIG. 13 is a graph of comparison plots of the uncommon port to uncommonport isolation of a 2 way lumped element power combiner/splitter havinga high pass configuration demonstrating the prior art isolationperformance and the advantages of one implementation of the presentinvention circuits, which has more uncommon port to uncommon portisolation over a wider bandwidth.

FIG. 14 illustrates a prior art circuit that is a 2-way lumped elementpower combiner/splitter using lumped elements having a low passconfiguration and a lumped element isolation resistor connected betweenuncommon ports.

FIG. 15 is a schematic circuit diagram of one embodiment of a 2-waymodified lumped element power combiner/splitter of the present inventionusing lumped elements having a low pass configuration and lumped elementisolation components in a lossy band response circuit configuration,connected between uncommon ports.

FIG. 16 is a graph showing a comparison plot of the uncommon port touncommon port isolation for a 2 way lumped element powercombiner/splitter using a low pass configuration demonstrating the priorart and the advantages of one implementation of the present inventioncircuits, which has more uncommon port to uncommon port isolation over awider bandwidth.

FIG. 17 illustrates a prior art circuit that is a 2-way Gysel powercombiner/splitter using distributed transmission lines and lumpedelement isolation resistors, connected between a point within thecircuit and circuit return or ground.

FIG. 18 is a schematic circuit diagram of one embodiment of a 2-waymodified Gysel power combiner/splitter of the present invention usingdistributed transmission lines and lumped element isolation componentsin a lossy band response circuit configuration connected between a pointwithin the circuit and circuit return or ground.

FIG. 19 is a graph showing a comparison plot of the uncommon port touncommon port isolation for a 2-way Gysel power combiner/splitterdemonstrating the prior art and the advantages of two implementations ofthe present invention circuits, which have more uncommon port touncommon port isolation over a wider bandwidth.

FIG. 20 illustrates a prior art circuit that is a 2-way ninety degreepower combiner/splitter hybrid using distributed transmission lines andlumped element isolation resistor, connected between a point in thecircuit and circuit return, or ground.

FIG. 21 is a schematic circuit diagram of one embodiment of a 2-waymodified ninety degree power combiner/splitter hybrid of the presentinvention using distributed transmission lines and lumped elementisolation components in a lossy band response circuit configurationconnected between a point in the circuit and circuit return or ground.

FIG. 22 is a comparison plot of the uncommon port to uncommon portisolation for a 2-way ninety degree power combiner/splitterdemonstrating the prior art and the advantages of two implementations ofthe present invention circuits, which have more uncommon port touncommon port isolation over a wider bandwidth.

FIG. 23 illustrates a prior art circuit that is a 2-way 180 degree powercombiner/splitter hybrid using distributed transmission lines and alumped element isolation resistor, connected between a point in thecircuit and circuit return or ground.

FIG. 24 is a schematic circuit diagram of one embodiment of a 2-waymodified 180 degree power combiner/splitter hybrid of the presentinvention using distributed transmission lines and lumped elementisolation components in a lossy band response configuration, connectedbetween strategic locations in the circuit and circuit return or ground.

FIG. 25 is a graph showing a comparison plot of the uncommon port touncommon port isolation for a 2-way 180 degree hybrid powercombiner/splitter demonstrating the prior art and the advantages of twoimplementations of the present invention circuits, which have moreuncommon port to uncommon port isolation over a wider bandwidth.

FIG. 26 illustrates a prior art circuit that is a lumped element couplerusing two lumped element transformers.

FIG. 27 is a schematic circuit diagram of one embodiment of a modifiedlumped element coupler of the present invention using two lumped elementtransformers and two lumped element resistors connected between portsthat are desired to be isolated.

FIG. 28 is a schematic circuit diagram of one embodiment of a modifiedlumped element coupler of the present invention using two lumped elementtransformers and two lumped element lossy band response circuits,connected between ports that are desired to be isolated.

FIG. 29 is a graph of comparison plots of the input port to isolatedport isolation for a lumped element coupler demonstrating the prior artand the advantages of two implementations of the present inventioncircuits, which have more isolation between the input ports and theisolated ports, over a wider bandwidth.

DETAILED DESCRIPTION

For purposes of this disclosure, the transmission line schematic symbolsthroughout all the figures represent both the signal carrying conductorand the return path or ground path conductor, such as is consistent forcoaxial cable, coplanar, stripline or microstripline transmission linesor other transmission line structures. Additionally, the schematicelements portrayed as capacitances, inductances and resistances may beconsidered to be lumped elements, and may be elements etched on acircuit board or may be separate parts connected to the circuit board,or some other construction, such as integrated circuit implementations.

For purposes of this disclosure “coupling” and “coupler” may refer toany circuit providing at least one path between a port and one or moreother ports either through a conductor, magnetic coupling, or electricfield coupling or some combination of the three.

For purposes of this disclosure an electrical isolation provided over a“greater” bandwidth relative to isolation provided by a passiveresistance in a prior art design is an electrical isolation that is overa bandwidth that is at least 10 percent wider at an isolation of atleast −20 dB. In some embodiments, an electrical isolation provided overa “greater” bandwidth relative to isolation provided by a passiveresistance in a prior art design is an electrical isolation that is overa bandwidth that is at least 10 percent wider at an isolation of atleast −30 dB. In other embodiments, an electrical isolation providedover a “greater” bandwidth relative to isolation provided by a passiveresistance in a prior art design is an electrical isolation that is overa bandwidth that is at least 10 percent wider at an isolation of atleast −50 dB.

For purposes of this disclosure, approximately 180 degrees is defined asa delta phase shift between a phase of a first signal and a phase of asecond signal that is equal to 180 degree+/−10 degrees or at least a 20dB reduction in a sum of the first and second signals.

For purposes of this disclosure, approximately equal amplitude isdefined as amplitudes differing by no more than 2 dB. Stateddifferently, approximately equal amplitude for first and second signalsmay be defined as a difference in amplitude between the first and secondsignals such that when a phase of a first signal and a phase of a secondsignal are 180 degree out of phase there will be at least a 20 dBreduction in a sum of the first and second signals relative to theamplitude of at least one of the first and second signals.

Referring to FIG. 1, a Wilkinson combiner/splitter includes common port1 and uncommon ports 2 and 3; each connected to a transmission line 4,5, and 6, respectively, having a characteristic impedance of Z ohms, andmay be electrically short. Transmission lines 5 and 6 are coupled orconnected to transmission line 4 by quarter wave transmission lines 7and 8 and by a resistor 9. Transmission lines 7 and 8 have acharacteristic impedance of Z*sqrt(3) ohms. Resistor 9 has a resistanceof Z*2 ohms.

In the embodiment of FIG. 2, a power combiner comprises threetransmission lines 13, 14, 15, which may have any electrical length, butmay be as short as possible. The transmission lines 13, 14, 15, have acharacteristic impedance Z ohms, which may be 50 Ohms. These three inputor output lines 13, 14, 15, which may have a similar impedance Z ohms,connect to the circuit and to ports 10, 11, 12, respectively. Thecircuit is also comprised of two transmission lines 16, 17, each havinga characteristic impedance that may be Z*sqrt(2) ohms, e.g. 70.7 Ohmsfor Z=50 ohms. The lengths of transmission lines 16, 17 may beapproximately a quarter wavelength at the design center frequency andjoin together to a common transmission line 13. The two uncommon ends ofthe transmission lines 16, 17 each connect to a different one of thetransmission lines 14, 15 and have a lossy band response circuitconnected between them.

In the illustrated embodiment, the lossy band response circuit includestwo capacitors 18, 22, a resistor 20, and two inductors 19, 21,hereinafter referred to as a ‘lossy band response circuit’. In general,a ‘lossy band response circuit’ may be defined as an amplitude and phasemodifying circuit comprised of any series resonant circuit, parallelresonant circuit, or some combination of the two types of circuits,single order or multiple order, that are comprised of lumped elementcomponents that incorporate some intentional resistance, i.e. aresistance that may include any parasitic resistance of the reactiveelements, but also includes additional added resistance that issubstantially greater than that parasitic resistance. Specifically, forpurposes of this disclosure a resistance that is substantially greaterthan a parasitic resistance is one that is at least 100 times greaterthan the parasitic resistance. For purposes of this disclosure a lossyband response circuit may be configured as a band pass circuit or as aband reject circuit or some combination of the two circuit types. Asused herein a lossy band response circuit is designed to be at a centerfrequency equal to, or approximately equal to (e.g. within 15%,preferably 5%), the center frequency and/or design frequency of thecircuit in which it is placed and may have useful operation over thesome portion of the bandwidth of the combiner. It may be that one lumpedcomponent might incorporate two or more of the reactances required toachieve the desired resonant circuit, (e.g. one inductor withintentional high losses may be used in lieu of a separate low lossinductor and a resistor) or that each reactance is individually achievedwith separate lumped element components.

The energy from a signal injected into port 11 will split into twosignals within the circuit, each having a portion of the original energyand will arrive at port 12 via two different paths. One path is throughtransmission lines 16 and 17, while the other path is through the lossyband response circuit (lumped elements 18 through 22). The following arethree possible scenarios that can occur with such a condition.

Scenario 1:

In the embodiment illustrated, the portion of the signal applied at port11 that goes through transmission lines 16 and 17, at the combinercenter frequency and then arrives at port 12, experiences a phase shiftof approximately one half a wavelength, or 180 degrees. The portion ofthe signal that goes through the lossy band response circuit (lumpedelements 18 through 22), has little or no phase shift due to the natureof the lossy band response circuit at the combiner center frequency. Theamplitudes of both signals that travelled different paths are nearly thesame. At port 12, the two portions of the original signal ‘add’together. Two summed signals of nearly equal amplitude but nearly 180degrees out of phase, sum to zero amplitude, or very close to zero.Therefore, at the center frequency of design, a signal injected intoport 11 will appear at port 12 having very small voltage amplitude. Thisis how high uncommon port to uncommon isolation is achieved. Thisexplanation, only at the center frequency of design, also applies to theprior-art Wilkinson power combiner shown in FIG. 1.

Scenario 2:

At frequencies slightly above (e.g. 1% or more) the combiner centerdesign frequency, the portion of the signal applied to port 11 travelingthrough transmission lines 16 and 17 experiences a phase shift greaterthan 180 degrees. However, the portion of the signal that travelsthrough the lossy band response circuit (lumped elements 18-22) receivesa phase shift greater than 0 degrees, making the delta phase shift stillclose to 180 degrees between portions of the signals that travel thedifferent paths but both arrive and sum at port 12. The amplitudes ofthe signals through each path are still nearly the same. Thus, a signalinjected into port 11 that is slightly above the center design frequencywill also nearly cancel at port 12, resulting in a relatively smallamplitude at port 12, which gives a relatively high uncommon port touncommon port isolation in the circuit.

Scenario 3

At frequencies slightly below (e.g. 1% or more) the combiner centerdesign frequency, the portion of the signal applied to port 11 andtraveling through transmission lines 16 and 17 experiences a phase shiftless than 180 degrees. The portion of the signal that travels throughthe lossy band response filter (lumped elements 18-22) receives a phaseshift less than 0 degrees, making the delta phase shift still close to180 degrees between portions of the signals that travel the differentpaths but both arrive and sum at port 12. The amplitudes of the signalsthrough each path are nearly the same. Thus, a signal injected into port11 that is slightly below the center design frequency will also nearlycancel, resulting in a relatively small voltage amplitude at port 12,which gives a relatively high uncommon port to uncommon port isolationin the circuit.

The exact values of the lumped elements 18-22 depend on what uncommonport to uncommon port isolation bandwidth is desired. For example, if itis desired to have an isolation of 50 dB over approximately a 20%bandwidth, for a design centered at 1 GHz designed for 50 ohms, theninductors 19 and 21 should have a reactance of approximately Z*1.08ohms, reactive, or 54.42 Ohms. Resistor 20 should be approximatelyZ*1.97 ohms or 98.79 Ohms, and capacitors 18 and 22 should beapproximately Z*1.08 ohms, reactive, or 54.16 Ohms. These values mayneed to be slightly adjusted for different designs, which will result inslightly different isolation values over the desired bandwidth. Otherimpedance combinations may be found as well to achieve 50 dB ofisolation over nearly a 20% bandwidth. If, on the other hand, it isdesired to achieve an isolation of 70 dB over a 5% bandwidth, then thefollowing impedances should be achieved. Inductors 19, 21 should have areactive impedance of Z*1.09 ohms, reactive, or 54.43 Ohms. Resistor 20should be Z*1.997 ohms, or 99.85 Ohms for, and capacitors 18, 22 shouldhave a reactive impedance of Z*1.09 ohms, reactive, or 54.41 Ohms, toachieve that isolation. Again, other impedances may be found thatachieve 70 dB of isolation over nearly a 5% bandwidth.

Note that the resistance of resistor 20 has a slightly different valuefrom the standard 100 Ohms that a Wilkinson power combiner requires in aZ=50 ohm design. This different resistance value from the prior artimplementation is common in the many implementations of the presentinvention. The resistance portion of the lossy band response circuit isdesigned or tuned to be a slightly different value than the standardvalue used in each of the prior art circuits, and may be slightly higherin resistance or more commonly, slightly lower in resistance. Inaddition, note that while the reactance impedances may vary a little fordifferent isolation bandwidths, the resistance change is a dominantfactor in realizing different isolation bandwidths.

For purposes of this disclosure the definition of “slightly” differentthan a nominal resistance value (100 Ohms, in the example above) may beinterpreted as different from the nominal value by between 0.01 percentand 25 percent. For the embodiments disclosed herein, increases in thedifference between the resistance portion of the lossy band responsecircuit and the nominal resistance value of an analogous conventionalcircuit will generally result in broadening of the isolation bandresponse of the combiner, coupler, or other device in which it isincorporated. However, the degree of isolation will generally decreasewith increasing the difference between the resistance of the resistanceportion and the nominal resistance value.

For a single section power combiner, the amplitude and phase matchingcircuit is defined as follows. It is a circuit that has nearly the sameamplitude response, in a desired bandwidth, from an uncommon port to anadjacent uncommon port, as the amplitude response the transmission linepath has between the same two uncommon ports. The amplitude and phasematching network also has a phase response that is nearly 180 degreesout of phase, in the same desired bandwidth, with the transmission linepath phase response. Therefore an amplitude and phase matched networkallows high isolation between uncommon ports to occur.

If the resistor 9 in FIG. 1 only is used to provide isolation from port2 to port 3, which is the configuration described by Wilkinson, thephase shift through the resistor 9 should be about 0 degrees,independent of frequency. Even though the signal amplitudes through thetwo paths (through transmission lines 7, 8 and the resistor path 9) maybe the same over some bandwidth, the phase difference is 180 degreesonly at one frequency, or over a small bandwidth, so that highisolations of greater than 50 dB can occur only over a small bandwidthof about 1%, centered at the design center frequency, as shown by line3A of FIG. 3.

The embodiment of FIG. 2 just described does not exhibit significantincrease in insertion loss over a typical Wilkinson combiner. The addedcomponents, including the inductors 19, 21 and capacitors 18, 22, may bemostly reactive with small resistive losses and do not dissipatesignificant energy.

FIG. 3 shows a graph of the combiner uncommon port to uncommon portisolation in dB from port 11 to port 12 in FIG. 2, for a 50 dB design aswell as a 70 dB design, both of the present invention, compared with thestandard isolation of a Wilkinson combiner (prior art), between uncommonports 2 and 3 as shown in FIG. 1. Line 3A is the uncommon port touncommon port isolation of a standard Wilkinson power combiner in dB.Line 3B is the uncommon port to port isolation in dB of animplementation of the embodiment of FIG. 2, showing approximately 50 dBisolation over nearly a 20% bandwidth. Line 3C is the uncommon port touncommon port isolation in dB of another implementation of the circuitof FIG. 2, showing approximately 70 dB isolation over nearly a 5%bandwidth.

The embodiment shown in FIG. 2 may be extended to include additionallumped element components, increasing the circuit order between ports 11and 12 to increase the isolation bandwidth as well as the operationalbandwidth of the power combiner/splitter.

The embodiment shown in FIG. 2 may also be modified to includeadditional quarter wave transmission line sections, such as is revealedby Cohn, to extend the bandwidth of the combiner/splitter even more.Each additional section would consist of two approximately quarter wavetransmission lines as Cohn has shown in the previous art, but replacingeach section's isolation resistor with a band response amplitude andphase matching circuit as described herein, i.e. a lossy band responsecircuit.

Referring to FIG. 4, for an M-section combiner/splitter, an amplitudeand phase matching circuit is defined as having the required phase andamplitude response to cause a signal injected into one uncommon port tobe canceled at another uncommon port. In an M-section combiner/splitter,M lossy band response circuits may be used, separated by approximatelyquarter-wave transmission line sections, (each section having impedancesas are required for an M section broadband Wilkinson combiner) and thatprovide high uncommon port to uncommon port isolation. With atwo-section power combiner, for example, 50 dB of isolation may beachieved over approximately a 100% (octave) bandwidth. 70 dB ofisolation may be achieved over approximately a 20% bandwidth.

It should be noted that all combiner/splitter designs may not requirethat the amplitude and phase matching circuit be symmetrical withrespect to each uncommon port between which it is connected, as shown inFIG. 2. Combiners designed for relatively low frequencies may use lossyband response circuits that are not symmetrical between uncommon ports,with little degradation of the output power(s). In addition, combinersthat do not require an even or equal split or combine ratio may also usean asymmetrical lossy band response circuit design such as one inductorin series with one resistor in series with one capacitor. Also, thearrangement of the components may not be important. For example,resistors could be connected directly to the uncommon ports rather thanthe capacitors, and the inductor could be in the middle of the lossyband response circuit.

As referenced earlier, FIG. 4 is a general diagram showing that it ispossible to extend the embodiments of the present invention to includemore than one section of approximately quarter wavelength transmissionlines. In doing so, the useful bandwidth of the power combiner canincrease and the uncommon port to uncommon port isolation can increaseover bandwidth. FIG. 4 also points out that the lossy band responsecircuits can be comprised of a simple RLC components, or can be made tobe a several order circuit, both with series components and shunt (tocircuit return) components.

FIG. 5 shows prior art of a three-way combiner/splitter. Input/outputports are common port 50 and uncommon ports 51, 52 and 53 that areconnected to transmission lines 54, 55, 56, 57, respectively, havingcharacteristic impedance Z ohms. Transmission lines 55, 56, 57 areconnected to transmission line 54 through quarter-wavelengthtransmission lines 58, 59, 60, respectively, each of the threetransmission lines having a characteristic impedance of Z*sqrt(3) ohms.Resistor 61 connects between transmission line 55 and transmission line56, resistor 62 connects between transmission line 56 and transmissionline 57, and resistor 63 connects between transmission line 57 andtransmission line 55. The resistors 61, 62 and 63 may have resistanceZ*3.08 ohms.

FIG. 6 shows an embodiment of a three-way power combiner/splitter inaccordance with an embodiment of the present invention, where again, theresistances 61, 62, 63 called for in the prior art circuit of FIG. 5,are replaced with lumped element lossy band response circuits, comprisedof components 81-83, 84-86, and 87-89, respectively, as shown in FIG. 6.

The circuit of FIG. 6 includes common port 70 and uncommon ports 71-73connected to transmission lines 74-77, respectively, havingcharacteristic impedance Z ohms. Transmission lines 75-77 are connectedto transmission line 74 through quarter-wavelength transmission lines78-80, respectively. The characteristic impedance of thequarter-wavelength transmission lines 78-80 may be Z*sqrt(3) ohms. In a50 ohm system, capacitors 81, 84 and 87 may be Z*3.6 ohms, reactive, or182.9 ohms. Inductors 83, 86 and 87 may be Z*3.8 ohms, reactive, or188.5 ohms. The values of the resistors 82, 85, 88 may be Z*2.84 ohms,or 142 Ohms.

FIG. 7 shows a graph of the combiner uncommon port to uncommon portisolation in dB from port 71 to port 72 in FIG. 6, for a 43 dB design(line 7B) as well as a 66 dB design (line 7C), both of the presentinvention, compared with the standard port to port isolation (line 7A)of a three-way prior art combiner, between uncommon ports 51 and 52 asshown in FIG. 5. Line 7A is the uncommon port to uncommon port isolationof a standard 3 way prior art power combiner in dB (see FIG. 5). Line 7Bis the uncommon port to uncommon port isolation in dB of the embodimentof FIG. 6, showing approximately 43 dB isolation over nearly a 32%bandwidth. Line 7C is the uncommon port to uncommon port isolation in dBof an embodiment of FIG. 6, showing approximately 66 dB isolation overnearly an 8% bandwidth.

Referring to FIG. 8, in this version of the prior-art Wilkinsoncombiner/splitter common port 90 and uncommon ports 91-93 are connectedto transmission lines 94-97, respectively, which have characteristicimpedance Z ohms. Transmission lines 95-97 are connected to transmissionline 94 by quarter-wavelength transmission lines 98-100, that may have acharacteristic impedance of Z*sqrt(3) ohms. Transmission lines 95-97 areconnected to a common node 104 through resistances 101-103,respectively. The resistances 101-103 may be Z ohms.

FIG. 9 shows an embodiment of the present design of a three-way powercombiner/splitter. The embodiment of FIG. 9 is a modified Wilkinsondesign, where, again, the resistances called for in the prior artcircuit of FIG. 8 (items 101, 102 and 103) are replaced with lumpedelement lossy band response circuits, comprised of components 121through 129 shown in FIG. 9.

In the embodiment of FIG. 9, common port 110 and uncommon ports 111-113are connected to transmission lines 114-117, respectively, which havecharacteristic impedance Z ohms. Transmission lines 115-117 areconnected to transmission line 114 by quarter-wavelength transmissionlines 118-120 that may have an impedance of Z*sqrt(3) ohms. Transmissionlines 115-117 are connected to a common node 130 through lossy bandresponse circuits. Specifically transmission lines 115-117 are connectedto common node 130 by elements 121-123, 124-126, 127-129, respectively.In a 50 ohm system, capacitors 122, 125 and 128 have a reactance ofapproximately Z*1.3 ohms, or 66 ohms. Inductors 123, 126 and 129 may beZ*1.3 ohms, or 66 ohms. Resistances 121, 124, 127 may be slightly more(e.g. 50.3 ohms) or less (e.g. 49.4 ohms) than impedance Z ohmsdepending on the desired isolation.

FIG. 10 shows a graph of the combiner uncommon port to uncommon portisolation in dB from port 111 to port 112 in FIG. 9, for a 45 dB design(line 10B) as well as a 67 dB design (line 10C), both of the presentinvention, compared with the standard port to port isolation (line 10A)of a three-way Wilkinson combiner (prior art), between uncommon ports 91and 92 as shown in FIG. 8. Line 10A is the uncommon port to uncommonport isolation of a standard 3 way Wilkinson power combiner (prior art)in dB (see FIG. 8). Line 10B is the uncommon port to uncommon portisolation in dB of the embodiment of FIG. 9, showing approximately 43 dBisolation over nearly a 32% bandwidth. Line 10C is the uncommon port touncommon port isolation in dB of the embodiment of FIG. 9, showingapproximately 66 dB isolation over nearly an 8% bandwidth.

In the prior art high-pass combiner of FIG. 11, common port 140 iscoupled or connected to uncommon ports 141, 142 by the illustratedcircuit. Port 141 is coupled to port 140 by capacitor 143 and acapacitor 144. Shunt inductor 147 connects an intermediate node betweencapacitors 143, 144 to ground 149. Port 142 is connected to port 140 bycapacitors 145, 146. An inductor 148 connects an intermediate nodebetween capacitors 145, 146 to ground 150. A resistor 151 connects ports141, 142 together. In a Z=50 ohm system, Capacitors 144 and 145 may beZ*1.28 ohms or 64 ohms, reactive. Capacitors 143 and 146 may beZ*sqrt(2) ohms or 70.7 ohms, reactive. Inductors 147 and 148 may beZ*sqrt(2) ohms or 70.7 ohms, reactive. Isolation resistor 151 may be Z*2ohms, or 100 ohms.

FIG. 12 shows an embodiment of a two-way power combiner/splitterincluding a solely lumped element high pass design, where, again, theresistance used in the prior art (e.g., resistor 151 of FIG. 11) isreplaced with a lumped element lossy band response circuit, comprised ofcomponents 171 through 177 shown in FIG. 12.

In the embodiment of FIG. 12, common port 160 is connected to uncommonports 161, 162 as shown in the illustrated circuit. Port 161 isconnected to port 160 by capacitors 163 and 164. Shunt inductor 167connects an intermediate node between capacitors 163, 164 to ground 169.Port 162 is connected to port 160 by capacitors 165 an 166. An inductor168 connects an intermediate node between capacitors 165, 166 to ground170. Ports 161, 162 are connected to one another by a lossy bandresponse circuit including, starting from port 161, a capacitor 171,inductor 172, resistor 173, inductor 174, and capacitor 175, in parallelwith an inductor 176 and a capacitor 177. In a 50 ohm system, capacitors171 and 175 may be Z*1.67 ohms or 83 ohms, reactive. Inductors 172 and174 may be Z*1.9 ohms, or 95 ohms, reactive. Inductor 176 may be Z*10.4or 519 ohms reactive. Capacitor 177 may be Z*4.6 or 230 ohms reactive.The value of resistor 173 may be Z*1.88 ohms or 94.1 ohms.

FIG. 13 shows a graph of the combiner uncommon port to uncommon portisolation in dB from port 161 to port 162 in FIG. 12, for a 57 dB design(line 13B) of the present invention, compared with the standard port toport isolation (line 13A) of a two-way solely lumped element high passcombiner (prior art), between uncommon ports 141, 142 as shown in FIG.11. Line 13A is the uncommon port to uncommon port isolation of theprior-art two-way solely lumped element high pass combiner of FIG. 11.Line 13B is the uncommon port to uncommon port isolation in dB of theembodiment of FIG. 12, showing approximately 57 dB isolation over nearlya 6% bandwidth.

In the prior-art high-pass combiner of FIG. 14, common port 180 isconnected to uncommon ports 181, 182 by the illustrated circuit. Port181 is connected to port 180 by an inductor 183. Shunt capacitor 185connects an intermediate node between inductor 183 and port 181 toground 188. Port 182 is connected to port 180 by inductor 184. A shuntcapacitor 187 connects an intermediate node between inductor 184 andport 182 to ground 190. A shunt capacitor 186 is also connected betweenport 180 and ground 189. A resistor 191 connects ports 181, 182 to oneanother. In a 50 ohm system, capacitor 186 may be Z/sqrt(2) ohms or 35ohms, reactive. Capacitors 185 and 187 may be Z/(2*sqrt(2)) ohms or 70.7ohms, reactive. Inductors 183 and 184 may be Z*sqrt(2) ohms or 70.7ohms, reactive. Isolation resistor 191 may be Z*2 ohms or 100 ohms.

FIG. 15 shows an embodiment of a two-way power combiner/splitterincluding a solely lumped element low pass design, often referred to asa lumped element Wilkinson design, where, again, the resistance used inthe prior art (see, e.g. resistor 191, FIG. 14) is replaced with alumped element lossy band response circuit, comprised of components212-218 as shown in FIG. 15.

In the embodiment of FIG. 15, common port 200 is connected to uncommonports 201, 202 as shown in the illustrated circuit. Port 201 isconnected to port 200 by an inductor 204. Shunt capacitor 206 connectsan intermediate node between inductor 204 and port 201 to ground 209.Port 202 is connected to port 200 by inductor 205. A shunt capacitor 208connects an intermediate node between inductor 205 and port 202 toground 211. A shunt capacitor 207 is also connected between port 200 andground 210. Ports 201, 202 are connected to one another by a lossy bandresponse circuit including, starting from port 201, a capacitor 212,inductor 213, resistor 214, inductor 215, and capacitor 216, in parallelwith an inductor 217 and a capacitor 218. In a 50 ohm system, capacitors212 and 216 may have an impedance of Z*2.3 ohms or 116 ohms, reactive.Inductors 213 and 215 may have an impedance of Z*2.5 ohms or 126 ohms,reactive. The value of resistor 214 may be Z*1.9 ohms, or 94.8 ohms.Inductor 217 may have an impedance of Z*2.1 or 104 ohms, reactive.Capacitor 218 may have an impedance of Z*1.7 or 86 ohms reactive.

FIG. 16 shows a graph of the combiner uncommon port to uncommon portisolation in dB from port 201 to port 202 for the embodiment of FIG. 15(line 16B) compared with the standard uncommon port to port isolation ofthe embodiment of FIG. 14 (line 16A). Line 16A is the uncommon port touncommon port isolation of the circuit of FIG. 14, from port 181 to 182.Line 16B is an uncommon port to uncommon port isolation in dB of theembodiment of FIG. 15 showing a high isolation design of approximately50 dB over nearly a 6% bandwidth.

FIG. 17 shows a prior art circuit 2 way combiner known as a Gyselcombiner. Port 220 is the common port and ports 221 and 222 are uncommonports. Transmission lines 223, 224 and 225 have characteristic impedanceZ, which may be 50 ohms and their length is typically electricallyshort. Transmission lines 226 and 227 may be Z*sqrt(2) ohm quarterwavelength transmission lines. Transmission lines 228 and 229 are Z ohmquarter wavelength transmission lines and join together at node 236through transmission lines 230 and 231. Transmission lines 230 and 231have impedance Z ohms and are quarter wavelength transmission lines atthe center frequency of design and join together at node 236. Resistors234 and 235 are both resistance Z ohms, and connect to transmissionlines 228 and 229, and 230, 231, respectively, and grounds 232 and 233.

FIG. 18 shows an embodiment of a two-way power combiner/splitterincluding a Gysel design, where, again, the isolation resistances usedin the prior art circuit of FIG. 17 (resistors 234 and 235) are replacedwith lumped element lossy band response circuits, comprised ofcomponents 254 through 259 shown in FIG. 18.

In FIG. 18, Port 240 is the common port and ports 241 and 242 areuncommon ports. Transmission lines 243, 244 and 245 have characteristicimpedance Z ohms, and their length is typically electrically short.Transmission lines 246 and 247 may be Z*sqrt(2) ohms and are a quarterwavelength at the center frequency of design. Transmission lines 248 and249 have impedance Z ohms, and are also a quarter wavelength.Transmission lines 250 and 251 have impedance Z ohms and are a quarterwavelength and join together at node 260. In a 50 ohm system, capacitors254 and 257 may be Z*1.1 ohms or 54.9 ohms, reactive. Inductors 256 and259 may be Z*1.12 ohms or 55.9 ohms, reactive, or 8.9 nH. Resistors 255and 258 may be Z*0.95 ohms, or 47.6 ohms. These values are for only oneembodiment of the present design. The lossy band response circuitsconnect to nodes between transmission lines 248 and 249, andtransmission lines 250, 251, and grounds 252 and 253 respectively.

FIG. 19 shows a graph of the combiner uncommon port to uncommon portisolation for a 36 dB implementation of the embodiment of FIG. 18 (line19B), a 45 dB implementation of the embodiment of FIG. 18 (line 19C),and the uncommon port to port isolation of the prior-art circuit of FIG.17 (line 19A). Line 19A is the uncommon port to uncommon port isolationof the two-way combiner splitter using the Gysel design as shown in FIG.17, between ports 221 and 222. Line 19B is the uncommon port to uncommonport isolation in dB, between ports 241 and 242 of an implementation ofthe embodiment of FIG. 18 showing approximately 36 dB isolation overnearly a 22% bandwidth. Line 19C is the uncommon port to uncommon portisolation in dB, between ports 241 and 242 of another implementation ofthe embodiment of FIG. 18 showing approximately 45 dB isolation overnearly a 14% bandwidth.

FIG. 20 shows a 90 degree hybrid circuit of prior art. Port 270 is thecommon port and ports 271 and 272 are uncommon ports. Transmission lines273, 274 and 275 have impedance Z and are typically short in electricallength. Transmission lines 276 and 277 may be Z*0.95 ohms and are onequarter wavelength at the center frequency of design. Transmission lines278 and 279 are Z ohms and are one quarter wavelength at the centerfrequency of design. Resistor 281 connects from a node betweentransmission line 277 and transmission line 278 to ground 280 and has avalue of Z ohms.

FIG. 21 shows an embodiment of a two-way power combiner/splitterincluding a 90 degree hybrid combiner design, where, again, theisolation resistor used in the prior art circuit of FIG. 20 (resistor281) is replaced with a lumped element lossy band response circuit,comprised of components 302 through 304 as shown in FIG. 21.

In FIG. 21, port 290 is the common port and ports 291 and 292 areuncommon ports. Transmission lines 294, 295 and 296 have impedance Zohms and are typically short in electrical length. Transmission lines297 and 298 may have impedance Z*0.71 ohms and may be one quarterwavelength at the center frequency of design. Transmission lines 299 and300 are impedance Z ohms and may be one quarter wavelength at the centerfrequency of design. The lossy band response circuit capacitor 302 mayhave a value of value of Z*3.54 ohms, the inductor 304 may have a valueof Z*3.51 ohms and resistor 303 may have a value of Z*0.94, for oneembodiment of the present invention. Components 302 through 304 connectbetween transmission lines 298 and 299, and ground 301.

FIG. 22 shows a graph of the combiner uncommon port to uncommon portisolation in dB from port 291 to port 292 in FIG. 21, for a 35 dB design(line 22B) as well as a 45 dB design (line 22C), compared with thestandard port to port isolation (line 22A) of a prior-art two-way 90degree hybrid combiner between uncommon ports 271 and 272 as shown inFIG. 20. Line 22A is the uncommon port to uncommon port isolation of thecircuit of FIG. 20. Line 22B is the uncommon port to uncommon portisolation of an implementation of the embodiment of FIG. 21 showingapproximately 35 dB isolation over nearly a 12% bandwidth. Line 22B isthe uncommon port to uncommon port isolation of another implementationof the embodiment of FIG. 21 showing approximately 45 dB isolation overnearly a 6% bandwidth.

FIG. 23 shows a 180 degree combiner designed in the prior art. Port 310is a common port. Ports 311 and 312 are uncommon ports. Transmissionlines 313, 314 and 315 are Z ohm transmission lines and are typicallyelectrically short. Transmission line 316 may have Z*sqrt(2) ohmsimpedance and is 270 degrees long at the center frequency of design.Transmission lines 317, 318 and 319 may have impedance Z*sqrt(2) ohmsand each have electrical length of 90 degrees at the center frequency ofdesign. Resistor 321 connects between transmission lines 319 and 318,and circuit ground 320, and may have a value of Z ohms.

FIG. 24 shows an embodiment of a two-way power combiner/splitterincluding a 180 degree hybrid design, where, again, the isolationresistance used in the prior art version of FIG. 23 (resistor 321), isreplaced with a lumped element lossy band response circuit, comprised ofcomponents 341 through 343 as shown in FIG. 24.

In FIG. 24, port 330 is a common port. Ports 331 and 332 are uncommonports. Transmission lines 333, 334 and 335 are impedance Z ohmtransmission lines and are typically electrically short. Transmissionline 336 may have Z*sqrt(2) ohms impedance and may be 270 degrees longat the center frequency of design. Transmission lines 337, 338 and 339may be Z*sqrt(2) ohms and each have electrical length of 90 degrees atthe center frequency of design. Resistor 342 connects betweentransmission lines 338 and 339, and circuit ground 340, and may have avalue of Z*1.05 ohms. Capacitor 341 connects between transmission lines338 and 339, and circuit ground 340 and may have a value of Z*1.03 ohms,reactive. Inductor 343 connects between transmission lines 338 and 339,and circuit ground 340, and may have a value of Z*1.03 ohms, reactive.

FIG. 25 shows a graph of the uncommon port to uncommon port isolation indB from port 331 to port 332 for a 38 dB (line 25B) and 48 dB (line 25C)implementation of the embodiment of FIG. 24 compared with the standardport to port isolation (line 25A) of a prior-art two-way 180 degreehybrid combiner as shown in FIG. 23. Line 25A is the uncommon port touncommon port isolation, between ports 311 and 312, of a standardtwo-way 180 degree hybrid power combiner as shown in FIG. 23. Line 25Bis the uncommon port to uncommon port isolation in dB of animplementation of the embodiment of FIG. 24 showing 38 dB of isolationover nearly a 16% bandwidth. Line 25C is the uncommon port to uncommonport isolation in dB of an implementation of the embodiment of FIG. 24showing approximately 48 dB isolation over nearly a 12% bandwidth.

FIG. 26 is a lumped element coupler as designed in the prior art. It maybe used in multiple ways. However, if port 350 is used as the inputport, port 351 could be the lowest loss output port. Port 352 could bethe coupled port and port 353 could be the isolated port. For a couplerdesign that exhibits a coupled port of 20 dB, and a center frequency of1 GHz, 50 ohms, transformers 354 and 355 could have primary inductancesof 567 nH, secondary inductances of 5.8 nH and coupling K=0.99. Ground356 is the circuit ground or return.

FIG. 27 shows an embodiment of a lumped element coupler design, whereisolation resistors 367, 368 are used to provide another path to theisolation ports, that is approximately 0 degrees phase shifted andnearly the same amplitude as the signal achieved through the originalcircuit. If port 360 is used as the input port, port 361 could be thelowest loss output port. Port 362 could be the coupled port and port 363could be the isolated port. For a coupler design that exhibits a coupledport of 20 dB, and a center frequency of 1 GHz, 50 ohms, transformers364 and 365 could have primary inductances of 567 nH, secondaryinductances of 5.8 nH and coupling K=0.99. Ground 366 is the circuitground. Isolation resistors 367 and 368 could have a value ofapproximately 97 kOhms.

FIG. 28 shows an embodiment of a lumped element coupler design, wherelossy band response circuits (items 377 through 382) are used to provideanother path to the isolation ports, that is approximately 0 degreesphase shifted and nearly the same amplitude as the signal achievedthrough the rest of the circuit. If port 370 is used as the input port,port 371 could be the lowest loss output port. Port 372 could be thecoupled port and port 373 could be the isolated port. For a couplerdesign that exhibits a coupled port of 20 dB, and a center frequency of1 GHz, 50 ohms, transformers 374 and 375 may have primary inductances of567 nH, secondary inductances of 5.8 nH and coupling K=0.99. Ground 376is the circuit ground. Isolation resistors 378 and 381 may have a valueof approximately 97 kOhms. Capacitors 377 and 380 may have a value of0.02 pF. Inductors 379 and 382 may have inductances of approximately11.2 mH.

FIG. 29 shows a graph (line 29B) of the isolation in dB from port 360 toport 363 for a high isolation design implementation of the embodiment ofFIG. 27. FIG. 29 also shows the isolation between ports 370 and 373 ofthe embodiment of FIG. 28 (line 29C) compared with the standard couplerisolation (line 29A) between ports 350 and 353 for the prior-art circuitin FIG. 26. Line 29A is the uncommon port to uncommon port isolation ofthe standard lumped element coupler of FIG. 26. Line 29B is the uncommonport to uncommon port isolation in dB of an implementation of theembodiment of FIG. 27 showing approximately 38 dB isolation over nearlya 16% bandwidth. Line 29C is the uncommon port to uncommon portisolation in dB of an implementation of the embodiment of FIG. 28showing approximately 48 dB isolation over nearly a 12% bandwidth.

While preferred embodiments of the present invention have beendescribed, it is to be understood that the embodiments described areonly illustrative and the scope of the invention is to be defined solelyby the appended claims when accorded a full range of equivalence, manyvariations and modifications naturally occurring to those skilled in theart from a pursual hereof.

What is claimed is:
 1. A coupler comprising: a mutual inductance couplerincluding— an input port an output port; a coupled port; an isolationport; a first inductor connected between the input port and the outputport; a second inductor connected between the coupled and the isolationport; a third inductor magnetically coupled to the first inductor andconnected between a ground potential and to isolation port; and a fourthinductor magnetically coupled to the second inductor and connectedbetween the ground potential and to the output port; and a lossy circuitconnected between the input port and the isolation port and having aresistance effective to receive an input signal from the input port andin response to the input signal output a first signal to the isolationport; wherein the mutual inductance coupler is effective to receive theinput signal from the input port and in response to the input signaloutput a second signal to the isolation port, the second signal havingapproximately equal amplitude and opposite phase to the first signal. 2.The coupler of claim 1, wherein the lossy circuit is a lossy bandresponse circuit.
 3. The coupler of claim 1, wherein the lossy circuitis at least one resistor.
 4. The coupler of claim 1, wherein: the lossycircuit is a first lossy circuit; a second lossy circuit is connectedbetween the coupled port and the output port and has a resistanceeffective to receive a coupled signal from the coupled port and inresponse to the coupled signal output a third signal to the output port;and the mutual inductance coupler is effective to receive the coupledsignal from the coupled port and in response to the coupled signaloutput a fourth signal to the output port, the fourth signal havingapproximately equal amplitude and opposite phase to the third signal.